1
Ferrite magnetic amplifier in flyback power supplies
For dual output flyback supplies that provide actual power (5V 2A and 12V 3A, both of which are ±5% regulated), zero load states are entered when the voltage reaches 12V and cannot be regulated within the 5% limit. Linear regulators are an enforceable solution, but they are still not ideal due to their high price and reduced efficiency.
Our recommended solution is to use a magnetic amplifier at the 12V output, even in flyback topologies. In order to reduce costs, it is recommended to use a ferrite magnetic amplifier. However, the control circuit of the ferrite magnetic amplifier is not used with the control circuit of the traditional rectangular hysteresis loop material (high permeability material). The ferrite's control circuitry (D1 and Q1) draws current to maintain power at the output. The circuit has been thoroughly tested. The transformer windings are designed for 5V and 13V outputs. The circuit achieves 12V output ± 5% regulation while achieving even less than 1W of input power (5V 300 mW and 12V zero load).
Figure 1
2
Overcurrent protection is provided using existing arc suppression circuitry
Consider 5V 2A and 12V 3A flyback power supplies. One of the key specifications of this power supply is to provide over-power protection (OPP) against the 5V output when the 12V output reaches no load or is extremely light. Both outputs require ± 5% voltage regulation.
For common solutions, the use of a sense resistor reduces cross-regulation and the fuse is expensive. There are now arc suppression circuits for overvoltage protection (OVP). The circuit can meet both OPP and regulation requirements, which can be achieved using a partial arc suppression circuit.
As can be seen from Figure 2, R1 and VR1 form an active false load at the 12V output, which allows 12V voltage regulation at light loads at the 12V output. When the 5V output is overloaded, the voltage on the 5V output will drop. False loads draw large amounts of current. The voltage drop across R1 can be used to sense this large amount of current. Q1 turns on and triggers the OPP circuit.
Figure 2
3
Active shunt regulator with false load
In the field of switching power supply products from in-line voltage AC to low-voltage DC, flyback is currently the most popular topology. One of the main reasons for this is its unique cost-effectiveness, which provides multiple output voltages by simply adding additional windings to the transformer secondary.
Typically, the feedback comes from the output, which has the strictest requirements for output tolerances. This output then defines the number of revolutions per volt for all other secondary windings. Due to the leakage inductance effect, the required output voltage cross-regulation cannot always be obtained at the output, especially if a given output may be unloaded or extremely lightly loaded due to the other outputs being fully loaded.
A back-up regulator or false load can be used to prevent the output voltage from rising in such situations. However, due to the increased cost and reduced efficiency caused by the increased cost and efficiency of the post-stage regulator or false load, they are not attractive enough, especially in recent years when regulatory requirements for no-load and/or standby input power consumption have become increasingly stringent in a variety of consumer applications, and this design is beginning to be left out in the cold. The active shunt regulator shown in Figure 3 not only solves the regulation problem, but also minimizes cost and efficiency impact.
Figure 3: Active Shunt Regulator for Multiple Output Flyback Converters.
The circuit works as follows: When both outputs are in regulator range, the resistor dividers R14 and R13 bias the triode Q5, leaving Q4 and Q1 off. Under such operating conditions, the current flowing through Q5 acts as a small false load at the 5V output.
The standard difference between the 5V output and the 3.3V output is 1.7V. When a load requires additional current from the 3.3V output and the load current output from the 5V output does not increase in equal amount, its output voltage increases compared to the voltage at the 3.3V output. With voltages varying by approximately more than 100 mV, Q5 biases off, turning on Q4 and Q1 and allowing current to flow from the 5V output to the 3.3V output. This current reduces the voltage at the 5V output, which in turn reduces the voltage difference between the two outputs.
The current flow in Q1 is determined by the voltage difference between the two outputs. As a result, the circuit allows both outputs to remain regulated independent of their load, even in the worst cases, such as a full load at the 3.3V output and no load at the 5V output. Q5 and Q4 in the design can provide temperature compensation because the VBE temperature changes in each transistor can cancel out each other. Diodes D8 and D9 are not required, but can be used to reduce power dissipation in Q1, eliminating the need to add heatsinks to the design.
The circuit reacts only to the relative difference between the two voltages and is essentially ineffective at full load and light load. Because the shunt regulator is connected from the 5V output to the 3.3V output, the active dissipation of the circuit can be reduced by 66% compared to a grounded shunt regulator. The result is high efficiency at full load and low power consumption from light load to no load.
4
High-voltage input switching power supply using StackFETs
Industrial equipment operating with three-phase alternating current often requires an auxiliary power stage that provides stable, low-voltage DC power to analog and digital circuits. Examples of such applications include industrial actuators, UPS systems, and energy meters.
The specifications of such power supplies are much stricter than those required for off-the-shelf standard switches. Not only are the input voltages higher in these applications, but devices designed for three-phase applications in industrial environments must also tolerate very wide fluctuations—including extended drop times, power surges, and accidental loss of one or more phases. Furthermore, the specified input voltage range for these auxiliary power supplies can be as wide as 57 VAC to 580 VAC.
Designing such a wide range of switching power supplies can be a challenge, mainly due to the high cost of high-voltage MOSFETs and the limitations of the dynamic range of traditional PWM control loops. StackFET technology allows the use of less expensive, 600V rated low-voltage MOSFETs and integrated power controllers from Power Integrations, allowing for simple, inexpensive switching power supplies that operate over a wide input voltage range.
Figure 4: Three-phase input 3W switching power supply with StackFET technology.
The circuit works as follows: The input current of the circuit can come from a three-phase three-wire or four-wire system, or even from a single-phase system. The three-phase rectifier consists of diodes D1-D8. Resistors R1-R4 provide inrush current limiting. If fusible resistors are used, they can be safely disconnected during a fault without the need for a separate fuse. The pi filter consists of C5, C6, C7, C8 and L1 and filters the rectified DC voltage.
Resistors R13 and R15 are used to balance the voltage between the input filter capacitors.
When the MOSFET in the integrated switch (U1) is turned on, the source of Q1 will be pulled low, R6, R7, and R8 will provide gate current, and the junction capacitance from VR1 to VR3 will be turned on Q1. Zener diode VR4 is used to limit the gate source voltage applied to Q1. When the MOSFET in U1 is turned off, the maximum drain voltage of U1 will be clamped by a 450 V clamp network consisting of VR1, VR2, and VR3. This limits the drain voltage of U1 to close to 450 V.
Any additional voltage at the end of the winding connected to Q1 is applied to Q1. This design efficiently distributes the total rectified input DC voltage and flyback voltage between Q1 and U1. Resistor R9 is used to limit high-frequency oscillations during switching, clamping networks VR5, D9, and R10 are used to limit peak voltages on the primary due to leakage during flyback intervals.
Output rectification is provided by D1. C2 is the output filter. L2 and C3 form a secondary filter to reduce switching ripple at the output.
When the output voltage exceeds the total voltage drop between the optocoupler diode and VR6, VR6 turns on. A change in the output voltage causes a change in the current flowing through the optocoupler diode in U2, which in turn changes the current flowing through the transistor in U2B. When this current exceeds the FB pin threshold current of U1, the next cycle is rejected. Output regulation can be achieved by controlling the number of enable and rejection periods. Once the switching cycle is turned on, it ends when the current rises to the internal current limit of U1. R11 is used to limit the current flowing through the optocoupler during transient loads and to adjust the gain of the feedback loop. Resistor R12 is used to bias the Zener diode VR6.
IC U1 (LNK 304) has built-in functions that protect the circuit from disappearing feedback signals, short circuits at the output, and overload. Since U1 is powered directly from its drain pin, there is no need to add additional bias windings to the transformer. C4 is used to provide internal power decoupling.
5
Choosing a good rectifier diode simplifies and reduces the cost of EMI filter circuitry in AC/DC converters
This circuit simplifies and reduces the cost of EMI filter circuits in AC/DC converters.
Emigration compliance with AC/DC power supplies requires the use of a large number of EMI filter devices, such as X capacitors and Y capacitors. The standard input circuitry for AC/DC power supplies includes a bridge rectifier for rectifying the input voltage (typically 50-60 Hz). Since this is a low frequency AC input voltage, standard diodes such as the 1N400X series diodes can be used, and another reason is that these diodes are the cheapest in price.
These filter devices are used to reduce the EMI generated by the power supply in order to comply with published EMI limits. However, because the measurements used to record EMI only begin at 150 kHz and the AC line voltage frequency is only 50 or 60 Hz, the standard diodes used in bridge rectifiers (see Figure 1) have a long reverse recovery time and are generally not directly related to EMI generation.
However, in the past, input filter circuits sometimes included capacitors in parallel with bridge rectifiers to suppress any high-frequency waveforms caused by low-frequency input voltage rectification.
If fast recovery diodes are used in bridge rectifiers, there is no need to use these capacitors. When the voltage between these diodes begins to reverse, they recover very quickly (see Figure 2). This reduces stray line inductance in the AC input line by reducing subsequent high-frequency shutdown spikes as well as EMI. Since 2 diodes can be turned on every half cycle, only 2 out of 4 diodes are of the fast recovery type. Similarly, in the two diodes that are turned on every half cycle, only one of the diodes needs to have fast recovery characteristics.
Figure 6: Typical input stage of SMPS using a bridge rectifier at the AC input.
Figure 7: The input voltage and current waveform shows the diode spurt at the end of reverse recovery.
6
Suppress low-cost output with soft-start to curb current spikes
To meet stringent standby power specifications, some multiple output power supplies are designed to disconnect the output when the standby signal is active.
Typically, this can be achieved by turning off a series bypass bipolar transistor (BJT) or MOSFET. For low current outputs, BJTs can be a suitable and inexpensive alternative to MOSFETs if the power transformer is designed with the additional voltage drop of the transistor in mind.
Figure 10 shows a simple BJT series bypass switch with a voltage of 12 V and an output current strength of 100 mA with a large capacitance (CLOAD). Transistor Q1 is a series bypass element that controls its switching based on the state of the standby signal. The value of resistor R1 is rated, which ensures that Q1 has enough base current to operate saturated at minimum Beta and maximum output currents. PI recommends adding an additional capacitor (Cnew) to regulate the transient current at on. If cnew is not added, Q1 quickly enters a capacitive load as soon as it is turned on, resulting in a large current spike. To regulate this transient spike, the capacity of Q1 needs to be increased, which leads to an increase in cost.
Cnew, used as an additional Q1 "Miller capacitor", eliminates current spikes. This additional capacitor limits the dv/dt value of the Q1 collector. The smaller the dv/dt value, the less charge current flows into the Cload. Specify a capacitance value for Cnew so that the ideal output value of Q1, dv/dt, multiplied by the Value of Cnew equals the current flowing into R1.
Figure 8: A simple soft-start circuit can disable power output during standby while eliminating current spikes at on,So small transistors (Q1) can be utilized to keep costs low.